Sensorless control of a brushless permanent-magnet motor

ABSTRACT

A method of controlling a brushless permanent-magnet motor. The method includes generating a first signal having a voltage that is proportional to a voltage across a winding of the motor, and generating a second signal having a voltage that is proportional to a current in the winding. The second signal is then differentiated to generate a third signal, and the voltages of the first signal and the third signal are compared. An output signal is generated in response to the comparison, the output signal having an edge whenever the voltages of the first signal and the third signal correspond. The winding is then commutated at times relative to the edges in the output signal. Additionally, a control system that implements the method, and a motor system that incorporates the control system.

REFERENCE TO RELATED APPLICATIONS

This application claims the priorities of United Kingdom Application No.1203911.1, filed Mar. 6, 2012, and United Kingdom Application No.1210371.9, filed Jun. 12, 2012, the entire contents of which areincorporated herein by reference.

FIELD OF THE INVENTION

The present invention relates to sensorless control of a brushlesspermanent-magnet motor.

BACKGROUND OF THE INVENTION

Knowledge of the rotor position is essential in order to commutate thephase windings of a brushless motor at the correct times. Apermanent-magnet motor will often include a Hall-effect sensor, whichoutputs a signal indicative of the rotor position. Although thecomponent cost of the sensor is relatively cheap, integrating the sensorwithin the motor often complicates the design and manufacture of themotor. Additionally, the signal output by the sensor is oftensusceptible to electromagnetic noise generated within the motor.

Sensorless schemes for determining indirectly the position of a rotorare known. For a permanent-magnet motor, transitions in the polarity ofthe back EMF induced in a phase winding may be used to determine therotor position. For a multi-phase motor, the rotor position may bedetermined by sensing the back EMF induced in a non-excited phasewinding. For a single-phase motor, the lack of additional phase windingsmakes this type of control unfeasible. Nevertheless, the position of therotor may be determined by suspending excitation at points in theelectrical cycle where transitions in the polarity of the back EMF areexpected. Unfortunately, suspending excitation has the disadvantage ofreducing the electrical power that can be driven into the motor.

SUMMARY OF THE INVENTION

In a first aspect, the present invention provides a method ofcontrolling a brushless permanent-magnet motor, the method comprisinggenerating a first signal having a voltage that is proportional to avoltage across a winding of the motor, generating a second signal havinga voltage that is proportional to a current in the winding,differentiating the second signal to generate a third signal, comparingthe voltages of the first signal and the third signal, generating anoutput signal in response to the comparison, wherein an edge isgenerated in the output signal when the voltages of the first signal andthe third signal correspond, and commutating the winding at timesrelative to edges in the output signal.

The rotor of a permanent-magnet motor induces a back EMF in the winding.The magnitude of the back EMF depends on, among other things, theangular position of the rotor. The voltage equation for winding may beexpressed as V_(ph)=i_(ph)R_(ph)+L_(ph)di_(ph)/dt+E_(ph), where V_(ph)is the voltage across the winding, i_(ph) is the current in the winding,R_(ph) is the resistance of the winding, L_(ph) is the inductance of thewinding, and E_(ph) is the back EMF induced in the winding by the rotor.For a motor having a relatively small i_(ph)R_(ph) term, the voltageequation reduces to V_(ph)=L_(ph)di_(ph)/dt+E_(ph). The voltage of thefirst signal is proportional to V_(ph), and the voltage of the thirdsignal is proportional to L_(ph)di_(ph)/dt. An edge is thereforegenerated in the output signal whenever the back EMF has a given value,which depends on how the voltages of the two signals are scaled. Sincethe magnitude of the back EMF depends on the angular position of therotor, an edge is generated in the output signal when the rotor is at apredetermined position. Accordingly, the position of the rotor may bedetermined without the need for a Hall-effect sensor.

The voltage equation holds true irrespective of the voltage across thewinding. Consequently, the position of the rotor may be determinedduring excitation of the winding. It is therefore possible to both drivethe motor and determine the position of the rotor using a singlewinding. The method does not require phase excitation to be suspended inorder to determine the rotor position. Accordingly, in comparison toexisting methods for sensorless control of a single-phase motor, moreelectrical power may be driven into the motor over each electricalhalf-cycle.

Zero-crossings in the back EMF occur when the rotor is at an aligned orunaligned position. At zero-crossings in the back EMF, the voltageequation further reduces to V_(ph)=L_(ph)di_(ph)/dt. The first and thirdvoltages of the first and third signals may therefore correspond inresponse to each zero-crossing in the back EMF. As a result, an edge isgenerated in the output signal whenever the rotor is at an aligned orunaligned position.

On commutating the winding, the direction of current through the windingis made to reverse. However, owing to the inductance of the winding,there is often a short period during which the current continues to flowin the same direction. Depending on how the current is sensed, thecurrent may appear to undergo an abrupt change in polarity in responseto commutation. As a result, a spike is created in the voltage of thethird signal. It is therefore possible that the voltages of the firstsignal and the third signal may correspond in response to commutatingthe winding. Accordingly, the method may comprise ignoring edges inresponse to commutating the winding.

The method may comprise exciting and freewheeling the winding. It maynot be possible to sense the current in the winding during freewheeling.As a result, the voltage of the second signal will be invalid, i.e. thevoltage will no longer be proportional to the magnitude of current inthe winding. This in turn may lead to spurious edges in the outputsignal. Accordingly, the method may comprise commutating the winding attimes relative to edges generated during excitation, and ignoring edgesgenerated during freewheeling. This then prevents the winding from beingcommutated at incorrect times during to spurious edges in the outputsignal.

It may be possible to measure the voltage across the winding and thecurrent in the winding during both excitation and freewheeling. That isto say that the method may comprise generating the first and secondsignals during both excitation and freewheeling. The method may thencomprise commutating the winding at times relative to edges generatedduring excitation or freewheeling. This then has the advantage that therotor position can be determined during either excitation orfreewheeling. As a result, advanced, synchronous or retarded commutationmay be employed.

The method may comprise exciting the winding with an excitation voltage,and generating a first signal having a voltage that is proportional tothe excitation voltage. As a consequence, the voltage of the firstsignal is proportional to the voltage across the winding duringexcitation only, i.e. the rotor position may be determined duringexcitation only. Nevertheless, this offers a relatively simple methodfor obtaining a measure of the voltage across the winding. Inparticular, a single potential divider may be used to generate the firstsignal.

In a second aspect, the present invention provides a control system fora brushless permanent-magnet motor, the control system performing amethod as claimed in any one of the preceding paragraphs.

In a third aspect, the present invention provides a control system for abrushless permanent-magnet motor, the control system comprising a firstsensor generating a first signal having a voltage that is proportionalto a voltage across a winding of the motor, a second sensor generating asecond signal having a voltage that is proportional to a current in thewinding, a differentiator differentiating the second signal andgenerating in response a third signal, a comparator comparing thevoltages of the first signal and the third signal and generating inresponse an output signal, wherein an edge is generated in the outputsignal when the voltages of the first signal and the third signalcorrespond, and a controller generating one or more control signals forcommutating the winding at times relative to edges in the output signal.

As noted above, the voltage equation for the winding may be expressed asV_(ph)=L_(ph)di_(ph)/dt+E_(ph). The voltage of the first signal isproportional to V_(ph), and the voltage of the third signal isproportional to L_(ph)di_(ph)/dt. An edge is therefore generated in theoutput signal whenever the back EMF has a given value, which depends onhow the voltages of the two signals are scaled. Since the magnitude ofthe back EMF depends on the angular position of the rotor, an edge isgenerated in the output signal when the rotor is at a predeterminedposition. Accordingly, the position of the rotor may be determinedwithout the need for a Hall-effect sensor.

The voltages of the first signal and the third signal may correspond inresponse to a zero-crossing in back EMF induced in the winding. An edgeis then generated in the output signal whenever the rotor is at analigned or unaligned position.

The first sensor and the second sensor may generate signals havingvoltages that are appropriately scaled such that the voltages of thefirst and third signals correspond at zero-crossings in the back EMF.Alternatively, scaling of at least one of the first signal, the secondsignal and the third signal may be required in order to ensure that thevoltages of the first and third signals correspond. Accordingly, thecontrol system may comprise a voltage scaler scaling at least one of thefirst signal, the second signal and the third signal such that thevoltages of the first signal and the third signal correspond in responseto a zero-crossing in back EMF induced in the winding. The voltagescaler may take the form of, for example, an amplifier or attenuator.

As noted above, it is possible that the voltages of the first signal andthe third signal will correspond in response to commutating the winding.Accordingly, the controller may ignores edges generated in response tocommutating the winding.

The controller may generate control signals for exciting andfreewheeling the winding. During freewheeling, it may not be possiblefor the second sensor to sense the current in the winding. Accordingly,the controller may generate control signals for commutating the windingin response to edges generated during excitation and ignores edgesgenerated during freewheeling. This then prevents the winding from beingcommutated at incorrect times during to spurious edges in the outputsignal.

The control system may comprise an inverter to which the winding iscoupled. The first sensor may then comprise a single potential dividerlocated across the inverter. As a consequence, the voltage of the firstsignal is proportional to the voltage across the winding duringexcitation only, i.e. the rotor position may be determined duringexcitation only. Nevertheless, a single potential divider offers arelatively simple and cost-effective means for obtaining a measure ofthe voltage across the winding. Alternatively, the first sensor maycomprise a pair of potential dividers located on opposite sides of thewinding, and the second sensor may comprise one of a current transducer,a current transformer and a pair of sense resistors located on oppositearms of the inverter. As a result, the voltage across the winding andthe current in the winding may be sensed during both excitation andfreewheeling. This then has the advantage that it is possible todetermine the rotor position during both excitation and freewheeling.Consequently, advanced, synchronous or retarded commutation may beemployed.

In a fourth aspect, the present invention provides a motor systemcomprising a brushless permanent-magnet motor and a control systemaccording to the preceding paragraph.

The motor may comprise a single phase winding. This then has theadvantage of simplifying the control system necessary to drive themotor. The control system is able to sense the position of the rotorduring excitation of the winding. Accordingly, sensorless control of asingle-phase motor may be achieved without compromising on electricalpower.

BRIEF DESCRIPTION OF THE DRAWINGS

In order that the present invention may be more readily understood,embodiments of the invention will now be described, by way of example,with reference to the accompanying drawings, in which:

FIG. 1 is a block diagram of a motor system in accordance with thepresent invention;

FIG. 2 is a schematic diagram of the motor system;

FIG. 3 details the allowed states of the inverter in response to controlsignals issued by the controller of the motor system;

FIG. 4 illustrates various waveforms of the motor system when operatingat a relatively low speed within acceleration mode;

FIG. 5 illustrates various waveforms of the motor system when operatingat a relatively high speed within acceleration mode;

FIG. 6 is a schematic diagram of the back EMF sensor of the motorsystem;

FIG. 7 illustrates various waveforms of the motor system when operatingwithin steady-state mode;

FIG. 8 is a schematic diagram of an alternative back EMF sensor of themotor system; and

FIG. 9 is a schematic diagram of an alternative motor system inaccordance with the present invention;

FIG. 10 illustrates various waveforms of the alternative motor systemwhen operating within steady-state mode;

FIG. 11 is a schematic diagram of a further alternative motor system inaccordance with the present invention;

FIG. 12 illustrates various waveforms of the further alternative motorsystem when operating within steady-state mode; and

FIG. 13 illustrates various waveforms of a still further motor system inaccordance with the present invention when operating at a relatively lowspeed within acceleration mode.

DETAILED DESCRIPTION OF THE INVENTION

The motor system 1 of FIGS. 1 and 2 is powered by a DC power supply 2and comprises a brushless motor 3 and a control system 4.

The motor 3 comprises a four-pole permanent-magnet rotor 5 that rotatesrelative to a four-pole stator 6. Conductive wires are wound about thestator 6 and are coupled together (e.g. in series or parallel) to form asingle phase winding 7.

The control system 4 comprises a DC link filter 8, an inverter 9, a gatedriver module 10, a current sensor 11, a back EMF sensor 12, and acontroller 13.

The DC link filter 8 comprises a capacitor C1 that smoothes therelatively high-frequency ripple that arises from switching of theinverter 9.

The inverter 9 comprises a full bridge of four power switches Q1-Q4 thatcouple the DC link voltage to the phase winding 7. Each of the switchesQ1-Q4 includes a freewheel diode.

The gate driver module 10 drives the opening and closing of the switchesQ1-Q4 in response to control signals received from the controller 13.

The current sensor 11 comprises a sense resistor R1 located on thenegative rail of the inverter 9. The voltage across the current sensor11 provides a measure of the current in the phase winding 7 whenconnected to the power supply 2. The voltage across the current sensor11 is output to the back EMF sensor 12 and the controller 13 as acurrent sense signal, I_SENSE.

The back EMF sensor 12 generates a digital signal, BEMF, which is outputto the controller 13. A more detailed description of the back EMF sensor12 is provided below in the section entitled Steady-State Mode.

The controller 13 comprises a microcontroller having a processor, amemory device, and a plurality of peripherals (e.g. ADC, comparators,timers etc.). The memory device stores instructions for execution by theprocessor, as well as control parameters (e.g. current limit, rise-timethreshold, speed threshold, freewheel period, advance period, conductionperiod, etc.) for use by the processor. The controller 13 is responsiblefor controlling the operation of the motor system 1 and generates threecontrol signals: DIR1, DIR2, and FW#. The control signals are output tothe gate driver module 10, which in response drives the opening andclosing of the switches Q1-Q4 of the inverter 9.

DIR1 and DIR2 control the direction of current through the inverter 9and thus through the phase winding 7. When DIR1 is pulled logically highand DIR2 is pulled logically low, the gate driver module 10 closesswitches Q1 and Q4, and opens switches Q2 and Q3, thus causing currentto be driven through the phase winding 7 from left to right. Conversely,when DIR2 is pulled logically high and DIR1 is pulled logically low, thegate driver module 10 closes switches Q2 and Q3, and opens switches Q1and Q4, thus causing current to be driven through the phase winding 7from right to left. Current in the phase winding 7 is thereforecommutated by reversing DIR1 and DIR2. If both DIR1 and DIR2 are pulledlogically low, the gate drive module 10 opens all switches Q1-Q4.

FW# is used to disconnect the phase winding 7 from the DC link voltageand allow current in the phase winding 7 to freewheel around thelow-side loop of the inverter 9. Accordingly, in response to a FW#signal that is pulled logically low, the gate driver module 10 causesboth high-side switches Q1,Q3 to open.

Each power switch Q1-Q4 conducts in a single direction only.Consequently, current freewheels through one of the low-side switchesQ2,Q4 and through a freewheel diode of the other low-side switch Q2,Q4.Certain types of power switch (e.g. MOSFETs) are capable of conductingin both directions. Accordingly, rather than freewheeling through afreewheel diode, both low-side switches Q2,Q4 may be closed such thatcurrent freewheels through both low-side switches Q2,Q4, i.e. inaddition to opening both high-side switches Q1,Q3, both low-sideswitches Q2,Q4 are closed in response to a logically low FW# signal.

FIG. 3 summarises the allowed states of the switches Q1-Q4 in responseto the control signals of the controller 13. Hereafter, the terms ‘set’and ‘clear’ will be used to indicate that a signal has been pulledlogically high and low respectively.

Excessive currents may damage components of the control system 4 (e.g.the power switches Q1-Q4) and/or demagnetise the rotor 5. The controller13 therefore monitors the current sense signal, I_SENSE, duringexcitation of the phase winding 7. In the event that current in thephase winding 7 exceeds a current limit, the controller 13 freewheelsthe phase winding 7 by clearing FW#. Freewheeling continues for afreewheel period, during which time current in the phase winding 7 fallsto a level below the current limit. At the end of the freewheel period,the controller 13 again excites the phase winding 7 by setting FW#. As aresult, current in the phase winding 7 is chopped at the current limit.

The controller 13 operates in one of three modes depending on the speedof the rotor 5. When the rotor 5 is stationary, the controller 13operates in start-up mode, which is employed merely to start the rotor 5moving in a forward direction. Once the rotor 5 is moving forwards, thecontroller 13 switches to acceleration mode. The controller 13 operatesin acceleration mode until the speed of the rotor 5 exceeds a speedthreshold, after which the controller 13 switches to steady-state mode.Within each mode of operation, the controller 13 employs a differentscheme to control the motor 3 without the need for a dedicated rotorsensor.

Start-Up Mode

The controller 13 makes no attempt to determine the position of therotor 5 when operating in start-up mode. Instead, the controller 13excites the phase winding 7 in a predetermined sequence that ensuresthat, irrespective of the position in which the rotor 5 has parked, therotor 5 is driven in a forwards direction.

The controller 13 begins by exciting the phase winding 7 in a particulardirection for a predetermined period of time. The choice of direction isunimportant. So, for example, the controller 13 might set DIR1 and clearDIR2 so as to excite the phase winding 7 from left to right.

The air gap between the poles of the stator 6 and the rotor 5 isasymmetric. As a result, the rotor 5 parks in a position for which therotor poles are misaligned slightly relative to the stator poles. Therotor 5 parks in one of two positions relative to the applied statorfield. In a first position, the rotor 5 is approximately aligned withthe applied stator field. In a second position, the rotor 5 isapproximately unaligned with the applied stator field. When parked inthe first position, the rotor 5 rotates backwards in response toexcitation of the phase winding 7. The rotor 5 rotates through arelatively small angle until the rotor 5 adopts the fully alignedposition. When parked in the second position, the rotor 5 rotatesforwards in response to excitation of the phase winding 7. The rotor 5rotates through a larger angle until the rotor 5 is again in the fullyaligned position. Accordingly, irrespective of the position at which therotor 5 has parked, the excitation of the phase winding 7 causes therotor 5 to move to the aligned position. The predetermined period overwhich the phase winding 7 is excited ensures that the rotor 5 moves fromeither parking position to the aligned position.

After exciting the phase winding 7 for the predetermined period, thecontroller 13 turns off the phase winding 7 by clearing both DIR1 andDIR2. By suspending phase excitation, the rotor 5 rotates forwardsthrough a small angle such that the rotor 5 adopts the first parkingposition. Phase excitation is suspended for a period sufficient toensure that the rotor 5 has come to a rest at the first parkingposition. The controller 13 then excites the phase winding 7 in theopposite direction to that previously employed. So, for example, thecontroller 13 might set DIR2 and clear DIR1 so as to excite the phasewinding 7 from right to left. This then causes the rotor 5 to be drivenforwards. At this stage, the controller 13 switches to accelerationmode.

Acceleration Mode

When operating in acceleration mode, the controller 13 employs a firstsensorless scheme for determining the position of the rotor 5.

On entering acceleration mode, the controller 13 is already exciting thephase winding 7. As noted above, the controller 13 employs acurrent-control scheme in which the phase winding 7 is freewheeledwhenever current in the winding 7 exceeds a current limit. Thecontroller 13 freewheels the phase winding 7 for a predeterminedfreewheel period. At the end of the freewheel period, the controller 13again excites the phase winding 7. The controller 13 thereforesequentially excites and freewheels the phase winding 7 over eachelectrical half-cycle.

The back EMF induced in the phase winding 7 influences the rate at whichcurrent in the phase winding 7 rises during excitation and falls duringfreewheeling. In particular, as the back EMF increases, current in thephase winding 7 rises at a slower rate and falls at a faster rate.Consequently, as the rotor 5 rotates, the phase current falls to adifferent level during each freewheel period. The phase currenttherefore starts from a different level and rises at a different rateduring each period of excitation. The applicant has found that the timetaken for the phase current to rise to the current limit during eachperiod of excitation depends on the angular position of the rotor 5.Moreover, the current-rise period decreases as the rotor 5 approaches analigned position. This finding is then exploited by the controller 13 inorder to determine the position of the rotor 5.

At the end of each freewheel period, the controller 13 starts a timer.When the phase current subsequently exceeds the current limit, thecontroller 13 stops the timer. The controller 13 then compares thecurrent-rise period stored by the timer against a predeterminedrise-time threshold. If the current-rise period is less than therise-time threshold, the controller 13 determines that the rotor 5 is atan aligned position.

When operating in acceleration mode, the controller 13 commutates thephase winding 7 in synchrony with each determined aligned position.Accordingly, in response to determining that the rotor 5 is at analigned position, the controller 13 immediately commutates the phasewinding 7 (i.e. by reversing DIR1 and DIR2, and setting FW#). However,if required, the controller 13 might alternatively commutate the phasewinding 7 before or after the determined aligned position. A scheme forcommutating the phase winding 7 at different times relative to thedetermined aligned position is described below in the section entitledSteady-State Mode.

The controller 13 assesses the position of the rotor 5 with each currentchop. Consequently, the frequency of current chopping defines theresolution with which the aligned position of the rotor 5 is determined.At relatively low rotor speeds, the length of each electrical half-cycleis relatively long and the magnitude of the back EMF is relativelysmall. As a result, the controller 13 typically chops the phase currentmany times over each electrical half-cycle and thus the aligned positionof the rotor 5 may be determined with relatively good accuracy. As thespeed of the rotor 5 increases, the length of each electrical half-cycledecreases and the magnitude of the back EMF increases. The controller 13therefore chops the phase current less frequently and thus the margin oferror in the aligned position determined by the controller 13 increases.By way of example, FIG. 4 illustrates waveforms for the phase current,the on/off signal for power switch Q1, and the back EMF over oneelectrical half-cycle when operating at a relatively low rotor speed.FIG. 5 then illustrates the same waveforms when operating at arelatively high rotor speed. It can be seen that the error in thedetermined aligned position (i.e. the difference between the actualaligned position and the aligned position determined by the controller13) is greater at the higher speed.

Owing to the behaviour identified in the previous paragraph, the valuechosen for the rise-time threshold influences the accuracy of thedetermined aligned position. For example, if the rise-time threshold isset too high, the controller 13 is likely to determine an alignedposition for the rotor 5 at an earlier point, particularly whenoperating at lower speeds. Conversely, if the rise-time threshold is settoo low, the controller 13 is likely to determine an aligned positionfor the rotor 5 at a later point, particularly when operating at higherspeeds. Since the phase winding 7 is commutated at a time relative tothe determined aligned position, the accuracy of the determined alignedposition influences the power and/or the efficiency of the motor system1.

In order to improve the accuracy of the determined aligned position, thecontroller 13 may employ a rise-time threshold that varies with rotorspeed. In particular, the controller 13 may employ a rise-time thresholdthat increases with increasing rotor speed. Consequently, at lowerspeeds, where the frequency of current chopping is relatively high, alower rise-time threshold may be used. Conversely, at higher speeds,where the frequency of current chopping is relatively low, a higherrise-time threshold may be used. As a result, the aligned position maybe determined with improved accuracy over a range of rotor speeds.

The accuracy of the determined aligned position may also be improved bydecreasing the freewheel period. As the freewheel period decreases, thefrequency of current chopping increases and thus the aligned positionmay be determined with improved accuracy. The minimum permissiblefreewheel period is likely to be dictated by the speed of the hardware(e.g. the speed of the controller 13 and the maximum switching frequencyof the power switches Q1-Q4) as well as the characteristics of the motor3 (e.g. the resistance and inductance of the phase winding 7, and theshape and magnitude of the back EMF).

Rather than employing a fixed freewheel period, it may be desirable tovary the freewheel period in response to changes in the rotor speed. Forexample, a longer freewheel period may be employed at lower speeds so asto minimise switching losses, and a shorter freewheel period may beemployed at higher speeds so as to increase the frequency of currentchopping and thus improve the accuracy of the aligned position.

The time spent in acceleration mode may be relatively short andtherefore the power and/or efficiency of the motor system 1 whenoperating in acceleration mode may be unimportant. Consequently, inspite of the advantages that arise when employing speed dependent valuesfor the rise-time threshold and/or the freewheel period, fixed valuesmay nevertheless be employed. Indeed, employing fixed values for therise-time threshold and/or freewheel period has the advantage ofsimplifying the control scheme.

The controller 13 determines the speed of the rotor 5 by measuring theinterval between two successive aligned positions, as determined by thecontroller 13. As noted above, there is a margin of error associatedwith each aligned position determined by the controller 13. Accordingly,in order to obtain a more accurate measure of the rotor speed, thecontroller 13 may measure the average interval for a plurality ofdetermined aligned positions. For example, the controller 13 maydetermine the rotor speed by averaging the interval over four successivealigned positions.

Immediately after the controller 13 has determined that the rotor 5 isat an aligned position, the position of the rotor 5 may still be at ornear the aligned position. Consequently, if the controller 13 were tocontinue measuring and comparing the current-rise period, furtheraligned positions may be determined for the same actual alignedposition. This is particularly true at lower rotor speeds, where themagnitude of the back EMF is relatively small and thus the rate at whichphase current rises during excitation is relatively fast. The controller13 therefore suspends measuring the current-rise period for apredetermined period of time after determining that the rotor 5 is atthe aligned position. This period will hereafter be referred to as thesuspension period. The suspension period is of a length that ensuresthat the rotor 5 is no longer at or near the aligned position at the endof suspension period. For example, the suspension period may be set suchthat, over the full speed range within acceleration mode, the rotor 5rotates through at least 70 electrical degrees during the suspensionperiod.

As the speed of the rotor 5 increases, the length of each electricalhalf-cycle decreases. Consequently, a fixed suspension period may beunsuitable, particularly when the speed range is relatively large. Forexample, the speed range when operating in acceleration mode may be1,000 to 50,000 rpm. At 1,000 rpm, the period of an electricalhalf-cycle for the four-pole motor is 15 ms. A suspension period of 5.8ms would therefore correspond to an electrical angle of about 70degrees. At 50,000 rpm, the period of an electrical half-cycle is 0.3ms. A suspension period of 5.8 ms is therefore clearly unsuitable atthis speed. Accordingly, the controller 13 may employ a suspensionperiod that varies with rotor speed. In particular, the controller 13may employ a suspension period that decreases with increasing rotorspeed.

In the scheme described above, the current-rise period starts at the endof the freewheel period. As a result, a single timer may be used forboth the current-rise period and the freewheel period. Nevertheless,since the freewheel period is predetermined, the current-rise periodmight alternatively start at the beginning of the freewheel period.Accordingly, in a more general sense, the current-rise period may besaid to be the interval between the start or end of freewheeling and thepoint in time at which current in the winding exceeds the current limit.

As the rotor speed increases, the period of each electrical half-cycledecreases and thus the time constant (L/R) associated with the phaseinductance becomes increasingly important. Additionally, the back EMFinduced in the phase winding 7 increases, which in turn influences therate at which current rises in the phase winding 7. It therefore becomesincreasingly difficult to drive current into the phase winding 7. Atrelatively high rotor speeds, the controller 13 may chop the phasecurrent just once or twice during each electrical half-cycle. As aresult, the margin of error in the aligned position determined by thecontroller 13 may be relatively large. The magnitude of the error mayadversely affect the power and/or efficiency of the motor system 1.Moreover, the error may prevent further acceleration of the motor 3.Accordingly, the controller 13 switches from acceleration mode tosteady-state mode when the speed of the rotor 5 exceeds a speedthreshold.

Steady-State Mode

When operating in steady-state mode, the controller 13 employs a secondsensorless scheme for determining the position of the rotor 5.

The second sensorless scheme makes use of the back EMF sensor 12. Aswill now be explained, the back EMF sensor 12 outputs a digital signalfor which certain edges correspond to aligned positions of the rotor 5.

In the absence of any significant saturation or saliency, the voltageequation for the phase winding 7 may be expressed as:

V _(ph) =i _(ph) R _(ph) +L _(ph) .di _(ph) /dt+E _(ph)

where V_(ph) is the voltage across the phase winding 7, i_(ph) is thecurrent in the phase winding 7, R_(ph) is the resistance of the phasewinding 7, L_(ph) is the inductance of the phase winding 7, and E_(ph)is the back EMF induced in the phase winding 7 by the rotor 5.

When the rotor 5 is at an aligned position, the back EMF induced in thephase winding 7 is zero. At each zero-crossing in the back EMF, thevoltage equation becomes:

V _(ph) =i _(ph) R _(ph) +L _(ph) .di _(ph) /dt

For reasons that are set out below, the i_(ph)R_(ph) term is negligiblearound zero-crossings in the back EMF. Consequently, for each alignedposition of the rotor 5, the voltage equation reduces to:

V _(ph) =L _(ph) .di _(ph) /dt

The back EMF sensor 12 makes use of this equation to generate an outputsignal having edges that correspond to aligned positions of the rotor 5.

As illustrated in FIG. 6, the back EMF sensor 12 comprises a voltagesensor 15, an amplifier 16, a differentiator 17, a low-pass filter 18,and a comparator 19.

The voltage sensor 15 comprises a potential divider R3,R4 that outputs afirst signal having a voltage that is proportional to the DC linkvoltage, V_(DC). When the phase winding 7 is excited, the voltage acrossthe phase winding, V_(ph), corresponds to the DC link voltage, V_(DC),minus the voltage drop across the power switches Q1-Q4. Consequently,the first signal output by the voltage sensor 15 has a voltage that isproportional to that across the phase winding, V_(ph), duringexcitation.

The amplifier 16 operates on the I_SENSE signal output by the currentsensor 11. The differentiator 17 then operates on the signal output ofthe amplifier 16, and the low-pass filter 18 operates on the signaloutput by the differentiator 17. The current sensor 11 may be said tooutput a second signal having a voltage that is proportional to currentin the phase winding 7. The differentiator 17 then differentiates thesecond signal and generates in response a third signal having a voltagethat is proportional to the rate of change of current in the phasewinding, di_(ph)/dt. The inductance of the phase winding 7 is assumed tobe constant (this is valid when the motor 3 has little or no saliencyand saturation effects are minimal) and thus the voltage of the thirdsignal is proportional to L_(ph).di_(ph)/dt.

The low-pass filter 18 is employed merely to suppress any noise that mayhave been introduced into the third signal by the differentiator 17. Ifnoise is not regarded as a problem then the filter 18 may be omitted. InFIG. 6, the low-pass filter 18 and the differentiator 17 are shown astwo distinct components. Alternatively, the low-pass filter 18 may beimplemented as part of the differentiator 17, thus avoiding the need foran additional operational amplifier.

The amplifier 16 ensures that the voltages of the first signal and thethird signal are scaled appropriately such that the voltages of the twosignals correspond when the back EMF induced in the phase winding 7 iszero, i.e. the voltages correspond when V_(ph)=L_(ph).di_(ph)/dt.Conceivably, the sense resistor R1 of the current sensor 11 and thepotential divider R3,R4 of the voltage sensor 15 may be configured suchthat the voltages of the first signal and the second signal are alreadyscaled appropriately, thereby avoiding the need for the amplifier 16.Alternatively, rather than an amplifier 16 operating on the secondsignal, a voltage scaler (e.g. amplifier or attenuator) may be used toscale one or more of the first signal, the second signal and the thirdsignal such that the voltages of the first signal and the third signalcorrespond when the back EMF is zero.

The comparator 19 compares the voltages of the first and third signalsand generates a digital output signal in response to the comparison. Theoutput signal is logically high (or alternatively logically low) whenthe voltage of the first signal is greater than the voltage of thesecond signal (i.e. when V_(ph)>L_(ph).di_(ph)/dt), and logically low(or alternatively logically high) when the voltage of the first signalis lower than the voltage of the second signal (i.e. whenV_(ph)<L_(ph).di_(ph)/dt). An edge is therefore generated in the outputsignal whenever the voltages of the two signals correspond, i.e.whenever V_(ph)=L_(ph).di_(ph)/dt. This condition is satisfied when theback EMF induced in the phase winding 7 is zero. Consequently, an edgeis generated in the output signal when the rotor 5 is at an alignedposition. However, as will now be explained, other edges are generatedin the output signal that do not correspond to aligned positions of therotor 5 and must therefore be ignored.

When operating in acceleration mode, the controller 13 sequentiallyexcites and freewheels the phase winding 7 over the full length of eachelectrical half-cycle. In contrast, when operating in steady-state mode,the controller 13 sequentially excites and freewheels the phase winding7 over a conduction period that spans only part of each electricalhalf-cycle. At the end of the conduction period, the controller 13freewheels the phase winding 7 by clearing FW#. Freewheeling thencontinues indefinitely until such time as the controller 13 commutatesthe phase winding 7. Within the region of falling back EMF, less torqueis achieved for a given phase current. Consequently, by freewheeling thephase winding 7 within this region, a more efficient motor system 1 maybe realised. Additionally, as the back EMF falls, the phase current mayrise sharply to an undesirable level. By freewheeling the phase winding7 in the region of falling back EMF, such current spikes may be avoided.

When operating in steady-state mode, the controller 13 commutates thephase winding 7 in advance of each aligned position of the rotor 5; thereasons for this are explained below. As noted in the previousparagraph, the phase winding 7 is freewheeling immediately prior tocommutation. During freewheeling, the phase current circulates aroundthe low-side loop of the inverter 9 and bypasses the current sensor 11.Consequently, no current passes through the current sensor 11 and thusthe voltage of the second signal is zero. In contrast, the currentflowing through the phase winding 7 is non-zero and may be relativelylarge. Accordingly, on commutating the phase winding 7, there is asudden change in the magnitude of the current through the current sensor11. Additionally, on commutating the phase winding 7, the polarity ofthe current through the current sensor 11 is initially negative, owingto the inductance of the phase winding 7. The current then rises sharplyand quickly becomes positive. Accordingly, on commutating the phasewinding 7, the voltage of the second signal output by the current sensor11 changes abruptly from zero to a negative value, and then risessharply to a positive value. As a result, the voltage of the thirdsignal output by the differentiator 17 has a negative spike (due to theabrupt change in the voltage of the second signal from zero to anegative value) immediately followed by a positive spike (due to thesharp rise in the voltage of the second signal). Owing to the positivespike in the voltage of the third signal, an edge is generated in theBEMF signal. However, this edge does not correspond to a zero-crossingin the back EMF. Instead, the edge is an artefact of the abrupt changein current through the current sensor 11, which arises because thecurrent sensor 11 is not able to sense the phase current duringfreewheeling.

As noted in the previous paragraph, current in the phase winding 7 risesrelatively quickly upon commutation. As a result, the voltage of thethird signal is relatively high and exceeds that of the first signal.However, as the back EMF decreases, crosses zero, and then opposes theDC link voltage, the rate of current rise decreases and thus the voltageof the third signal decreases. At some point, the voltage of the thirdsignal corresponds to that of the first signal and a further edge isgenerated in the BEMF signal. This edge then corresponds to azero-crossing in the back EMF.

Consequently, on commutating the phase winding 7, a first edge isgenerated in the BEMF signal due to the abrupt change in current throughthe current sensor 11. This is then followed by a second edge in theBEMF signal due to the zero-crossing in the back EMF. The controller 13therefore ignores the first edge and determines that the rotor 5 is atan aligned position in response to the second edge.

In response to detecting the second edge, the controller 13 ignores theBEMF signal until such time as the controller 13 again commutates thephase winding 7. The reasons for this are as follows. As in accelerationmode, the controller 13 freewheels the phase winding 7 for apredetermined freewheel period whenever current in the phase winding 7exceeds the current limit. Accordingly, during each conduction period,the controller 13 may chop the phase current. Should current choppingoccur, the current through the current sensor 11 will change abruptly asthe controller 13 freewheels and then excites the phase winding 7. Anyabrupt changes in the current through the sensor 11 may result inspurious edges in the BEMF signal. Additionally, during any freewheelperiod, the phase current circulates around the low-side loop of theinverter 9 and bypasses the current sensor 11. Consequently, the voltageof the second signal output by the current sensor 11 is zero, and thusthe third signal output by the differentiator 17 is invalid. Thecontroller 13 therefore begins to monitor the BEMF signal only inresponse to commutating the phase winding 7. The controller 13 thenignores the first edge in the BEMF signal and determines that the rotor5 is at an aligned position in response to the second edge in the BEMFsignal. Thereafter, the controller 13 ignores the BEMF signal until suchtime as the controller 13 again commutates the phase winding 7. In thepresent embodiment, the first edge is a rising edge and the second edgeis a falling edge. The controller 13 therefore monitors the BEMF signaland determines that the rotor 5 is at an aligned position in response toa falling edge.

In an attempt to demonstrate the behaviour of the motor system 1, FIG. 7illustrates possible waveforms for the phase current, the voltage of thesecond signal, the voltage of the third signal, and the BEMF signal overone electrical cycle. On commutating the phase winding 7, the voltage ofthe second signal can be seen to change abruptly from zero to a negativevalue, and then rise sharply to a positive value. As a result, thevoltage of the third signal has a negative spike (corresponding to theabrupt change in the voltage of the second signal from zero to anegative value) immediately followed by a positive spike (correspondingto the sharp rise in the voltage of the second signal). The magnitude ofthe positive spike is such that the voltage of the third signal exceedsthat of the first signal and thus a rising edge is generated in the BEMFsignal. As the voltage of the third signal decreases, the voltages ofthe first signal and the third signal again correspond and a fallingedge is generated in the BEMF signal.

At noted above, the controller 13 commutates the phase winding 7 inadvance of each aligned position of the rotor 5. The reasons for thisare as follows. During excitation, the phase voltage, V_(ph), isproportional to the DC link voltage, V_(Dc). During freewheeling, on theother hand, the phase voltage is zero. Consequently, the voltage of thesignal output by the voltage sensor 15 is proportional to the phasevoltage, V_(ph), only when the phase winding 7 is excited. Additionally,the current sensor 11 provides a measure of the phase current duringexcitation only. The back EMF sensor 12 is therefore capable ofdetermining the rotor position only when the phase winding 7 is excited.The controller 13 therefore commutates the phase winding 7 in advance ofeach aligned position of the rotor 5. This then ensures that the phasewinding 7 is excited as the rotor 5 passes through each alignedposition.

In order to commutate the phase winding 7, the controller 13 acts inresponse to each falling edge of the BEMF signal. In response to afalling edge of the BEMF signal, the controller 13 subtracts an advanceperiod, T_ADV, from the period of the electrical half-cycle, T_HC, inorder to obtain a commutation period, T_COM:

T_COM=T_HC−T_ADV

The controller 13 then commutates the phase winding 7 at a time, T_COM,after the falling edge. As a result, the controller 13 commutates thephase winding 7 in advance of the next aligned position of the rotor 5by the advance period, T_ADV. The period of the electrical half-cycle,T_HC, is defined by the interval between two successive edges of theBEMF signal.

The advance period defines the phase of excitation (i.e. the angle atwhich the phase winding 7 is excited relative to aligned positions ofthe rotor 5) and the conduction period defines the length of excitation(i.e. the angle over which the phase winding 7 is excited). Thecontroller 13 may adjust the advance period and/or the conduction periodin response to changes in the speed of the rotor 5. For example, thecontroller 13 may adjust the advance period and/or the conduction periodsuch that the same input or output power is achieved over a range ofrotor speeds.

In the discussion above, it was noted that the resistive term,i_(ph)R_(ph), of the phase voltage equation is negligible aroundzero-crossings in the back EMF. There are several reasons for this.First, the resistive term is relatively small irrespective of the rotorposition. For example, the phase resistance at 70 degrees C. may be0.03Ω, the voltage of the power supply 2 may be 24 V, and the currentlimit may be 30 A. Accordingly, when the phase current is at a maximumof 30 A, the i_(ph)R_(ph) term is 0.9 V. The phase voltage on the otherhand, is around 24 V. Accordingly, irrespective of the rotor position,the voltage equation for the phase winding 7 is dominated by theinductive and back EMF terms. Second, when operating in steady-statemode, the controller 13 freewheels the phase winding 7 during a periodof falling back EMF. Consequently, the phase current has been decayingfor a period prior to the zero-crossing in back EMF. Third, thecontroller 13 commutates the phase winding 7 in advance ofzero-crossings in the back EMF. Commutation naturally involves azero-crossing in the phase current. Since the phase current is generallynon-zero prior to commutation, and commutation occurs before azero-crossing in the back EMF, the zero-crossing in the phase currentwill occur at or near the zero-crossing in the back EMF. Consequently,the resistive term is negligible at zero-crossings in the back EMF.

At the speed threshold, the frequency of current chopping withinacceleration mode may be relatively low. As a result, the margin oferror in the aligned position determined by the controller 13 may berelatively large. Steady-state mode requires that the phase winding 7 isexcited in advance of rotor aligned positions. However, if the error inthe determined aligned position is relatively large, it is possible thatcommutation may occur at or after the rotor aligned position.Accordingly, prior to entering steady-state mode it may be necessary ordesirable to establish an aligned position for the rotor 5 with improvedaccuracy. Accordingly, as illustrated in FIG. 8, the back EMF sensor 12may comprise a pair of potential dividers R5,R6 and R7,R8, adifferential amplifier 20, and a zero-cross detector 21. The outputs ofthe potential dividers R5,R6,R7,R8 are fed to the amplifier 20, whichoutputs a measure of the phase voltage. The output of the amplifier 20is fed to the zero-crossing detector 21, which outputs a digital signalhaving edges that correspond to zero-crossings in the phase voltage. Ontransitioning from acceleration mode to steady-state mode, thecontroller 13 opens all switches Q1-Q4 of the inverter 9 and monitorsthe signal output by the zero-cross detector 21. With all switches open,the phase current decays through the freewheel diodes of the switchesQ1-Q4 until no current flows through the phase winding 7. At this point,the voltage across the phase winding 7 corresponds to the back EMFinduced in the phase winding 7. Consequently, each edge of the signaloutput by the zero-cross detector 21 corresponds to a zero-crossing inthe back EMF. The controller 13 therefore obtains a more accuratemeasure of the rotor aligned position. After a rotor aligned positionhas been identified (i.e. after an edge in the signal output by thezero-cross detector 21 has been sensed), the controller 13 switches tosteady-state mode and controls the motor 3 in the manner describedabove. In particular, the controller 13 monitors the BEMF signal andcommutates the phase winding 7 in response to falling edges in the BEMFsignal.

In the embodiment described above, the back EMF sensor 12 is capable ofsensing zero-crossings of back EMF during phase excitation only. Thecontroller 13 is therefore required to commutate the phase winding 7 inadvance of each zero-crossing in the back EMF. This is not regarded as aproblem since, at the relatively high speeds that occur withinsteady-state mode, advanced commutation is generally required in orderto drive sufficient current and thus power into the phase winding 7 overeach electrical half-cycle. Nevertheless, there may be instances forwhich it is desirable to synchronise or retard commutation relative tothe zero-crossings of back EMF.

Advanced, synchronised and retarded commutation may all be achievedusing the same control scheme as that described above. In response to afalling edge in the BEMF signal, the controller 13 subtracts a phaseperiod, T_PHASE, from the period of the electrical half-cycle, T_HC, inorder to obtain the commutation period, T_COM:

T_COM=T_HC−T_PHASE

The controller 13 then commutates the phase winding 7 at a time, T_COM,after the falling edge. As a result, the controller 13 commutates thephase winding 7 relative to the next rotor aligned position by the phaseperiod, T_PHASE. If the phase period is positive, commutation occursbefore the rotor aligned position (advanced commutation). If the phaseperiod is zero, commutation occurs at the rotor aligned position(synchronous commutation). And if the phase period is negative,commutation occurs after the rotor aligned position (retardedcommutation).

If synchronised or retarded commutation is employed, rotor alignedpositions are likely to occur when the phase winding 7 is freewheeling.An alternative design of current sensor 11 and voltage sensor 15 willtherefore be required in order to measure the phase voltage and thephase current during freewheeling as well as excitation.

FIG. 9 illustrates an alternative embodiment in which the current sensor11 and the voltage sensor 15 are different. In all other respects, thecontrol system 4 is unchanged. In particular, the back EMF sensor 12continues to comprise an amplifier 16, a differentiator 17, a low-passfilter 18, and a comparator 19.

The current sensor 11 comprises a pair of sense resistors R1 and R2, anda multiplexer 25. Each resistor R1,R2 is located on a lower leg of theinverter 9, with one of the resistors R2 providing a measure of thephase current when excited from left to right, and the other resistor R1providing a measure of the phase current when excited from right toleft. The multiplexer 25 selects one of the two signals output by thesense resistors R1,R2.

The voltage sensor 15 comprises a pair of potential dividers R5,R6 andR7,R8, a pair of differential amplifiers 22,23 and a multiplexer 24. Thepotential dividers R5,R6 and R7,R8 are located on opposite sides of thephase winding 7, and the outputs of the dividers R5,R6,R7,R8 are fed toboth differential amplifiers 22,23. The signal output by one of theamplifiers 22 provides a measure of the phase voltage when excited fromleft to right, and the signal output by the other amplifier 23 providesa measure of the phase voltage when excited from right to left. Themultiplexer 24 selects one of the two signals output by the amplifiers22,23.

The DIR1 signal output by the controller 13 is used as the selectorinput for both multiplexers 24,25. Accordingly, the multiplexers 24,25select one of the amplifiers 22,23 and one of the sense resistors R1,R2according to the direction of current through the phase winding 7. Bylocating the potential dividers R5,R6,R7,R8 on opposite sides of thephase winding 7, and by locating the sense resistors R1,R2 on oppositelegs of the inverter 9, the phase voltage and the phase current may besensed during freewheeling as well as excitation.

The back EMF sensor 12 of FIG. 6 is not able to sense the phase currentor phase voltage during freewheeling. As a result, spurious edges aregenerated in the BEMF signal during freewheeling. The back EMF sensor 12of FIG. 9, on the other hand, is able to sense the phase current and thephase voltage during both excitation and freewheeling. As a result, nospurious edges are generated during freewheeling. Nevertheless, a risingedge continues to be generated in the BEMF signal on commutating thephase winding 7. The rising edge occurs because the current through theselected sense resistor R1 or R2 is initially negative, owing to theinductance of the phase winding. As a result, a negative and thenpositive spike occurs in the voltage of the third signal. The controller13 therefore monitors the BEMF signal during excitation and freewheelingand commutates the phase winding 7 at times relative to the fallingedges of the BEMF signal.

FIG. 10 illustrates waveforms for the phase current, the voltage of thesecond signal, the voltage of the third signal, and the BEMF signal overone electrical cycle when employing the arrangement of FIG. 9.

FIG. 11 illustrates a further alternative embodiment in which thecurrent sensor 11 and the voltage sensor 15 are again changed.

The current sensor 11 comprises a current transformer 26 that senses thephase current during both excitation and freewheeling. The polarity ofthe signal output by the current transformer 26 reflects the directionof current through the phase winding 7.

The voltage sensor 15 comprises a pair of potential dividers R5,R6 andR7,R8 located on opposite sides of the phase winding 7, the outputs ofwhich are fed to a single differential amplifier 22. The signal outputby the amplifier 22 provides a measure of the phase voltage, with thepolarity of the voltage reflecting the direction of excitation, i.e. thephase voltage is positive when the phase winding 7 is excited from leftto right, and negative when excited from right to left.

As with the embodiment illustrated in FIG. 9, the current sensor 11 andthe voltage sensor 15 sense the phase current and the phase voltageduring both excitation and freewheeling. However, in contrast to theembodiment of FIG. 9, the voltage of the signal output by the currentsensor 11 does not undergo an abrupt change on commutating the phasewinding 7. As will now be explained, this has important implications forthe BEMF signal output by the back EMF sensor 12.

FIG. 12 illustrates waveforms for the phase current, the voltage of thesecond signal, the voltage of the third signal, and the BEMF signal overone electrical cycle when employing the arrangement of FIG. 11. It canbe seen that the voltage of the second signal output by the currentsensor 11 mirrors that of the phase current. In contrast to waveformillustrated in FIG. 10, there are no abrupt changes in the voltage ofthe second signal at the point of commutation. Consequently, there areno negative spikes in the voltage of the third signal. It may appearfrom FIG. 12 that the voltages of the first and third signals correspondat the point of freewheeling and at the point of commutation. However,this is not the case. Instead, the voltages of the two signals rise andfall together. Accordingly, the voltages of the two signals do notcorrespond and no edges are generated in the back EMF signal. Indeed, asis evident from FIG. 12, edges are generated in the BEMF signal only ata zero-crossing in the back EMF, i.e. when V_(ph)=L_(ph).di_(ph)/dt. TheBEMF signal therefore resembles the signal output by a conventionalHall-effect sensor.

Particular embodiments have thus far been described for measuring thephase voltage and the phase current. It will be appreciated that otherarrangements exist for measuring the voltage and current. By way ofexample only, the current sensor 11 may comprise a Hall-effect sensor orother current transducer.

In the embodiments described above, the back EMF sensor 12 is distinctfrom the controller 13 and is implemented in hardware external to thecontroller 13. Conceivably, however, if the required hardware forms partof the peripherals of the controller 13, one or more components of theback EMF sensor 12 (e.g. the amplifier 16, the differentiator 17, thelow-pass filter 18, and/or the comparator 19) may form an integral partof the controller 13.

The low-pass filter 18 of the back EMF sensor 12 may introduce a phasedelay into the third signal, which would in turn lead to a phase shiftin the BEMF signal. The filter 18 is therefore configured to removesufficient noise from the third signal with as little phase delay aspossible.

Two schemes have thus far been described for sensing the position of therotor 5. In the first scheme (employed in acceleration mode), the timetaken for the phase current to exceed a current limit is used todetermine the position of the rotor 5. In the second scheme (employed insteady-state mode), a comparison of the phase voltage and the rate ofchange of phase current is made in order to determine the position ofthe rotor 5.

The first scheme has the advantage that it can be implemented withoutthe need for any additional hardware. Indeed, in comparison to aconventional motor system that employs a Hall-effect sensor, the firstscheme employs at least one fewer component. When the phase currentexceeds the current limit, the phase winding 7 is freewheeled for afreewheel period. It is not therefore necessary to sense the phasecurrent during freewheeling. As a result, the first scheme may beimplemented using a single sense resistor to measure the phase current.The first scheme therefore offers a cost-effective method of determiningthe rotor position.

The first scheme relies on current chopping in order to determine theposition of the rotor 5. Moreover, the frequency of current choppingdetermines the resolution and thus the accuracy with which the rotorposition is determined. Consequently, when current chopping isrelatively infrequent (e.g. at relatively high speeds), the accuracy ofthe determined rotor position may be relatively poor. The second schemehas the advantage that it does not rely on current chopping in order todetermine the position of the rotor 5. Consequently, a determination ofthe rotor position may be made irrespective of rotor speed. The accuracyof the determined rotor position is defined partly by the resistiveterm, i_(ph)R_(ph). Fortunately, the resistive term is typically smalland can often be neglected. Moreover, even if the resistive term may besaid to be significant, the term simply increases the error in thedetermined aligned position. It nevertheless continues to be possible todetermine an aligned position for the rotor. A disadvantage with thesecond scheme lies in the additional hardware that is required toimplement the scheme, which naturally increases the cost of the motorsystem 1. Nevertheless, the increase in component cost may be offset bythe reduction in assembly cost as a result of omitting the Hall-effectsensor.

The cost of implementing the second scheme can be kept relatively low byemploying a single potential divider R3,R4 for the voltage sensor 15 anda single sense resistor R1 for the current sensor 11, as illustrated inFIG. 6. In employing a single potential divider R3,R4 and single senseresistor R1, the position of the rotor 5 can be sensed only when thephase winding 7 is excited. This is achieved by commutating the phasewinding 7 in advance of rotor aligned positions. At relatively lowspeeds, the phase current rises relatively quickly to the current limit.It is therefore possible that a zero-crossing in the back EMF may occurat a time when the phase winding 7 is freewheeling. Accordingly, thisparticular implementation of the second scheme, whilst cost-effective,may be unsuitable at relatively low speeds. However, by employing thefirst scheme at lower speeds and the second scheme at higher speeds, acost-effective solution is obtained for controlling the motor 3 over thefull range of rotor speeds.

The power supply 2 may output a voltage that varies with time. Forexample, the power supply 2 may comprise a battery that discharges withuse. Alternatively, the power supply 2 may comprise an AC source andrectifier that provide a rectified voltage. Depending on the capacitanceof the DC link filter 8, the DC link voltage may have a relatively highripple. Alternatively, the DC link filter 8 may smooth the rectifiedvoltage but the RMS voltage of the AC source may drift with time. Therate at which current rises in the phase winding 7 depends on themagnitude of the phase voltage. Accordingly, when employing the firstsensorless scheme, any changes in the voltage of the power supply 2 mayaffect the point at which an aligned position is determined by thecontroller 13. For example, as the voltage of the power supply 2increases, the rate at which phase current rises increases and thus thelength of the current-rise period decreases. If the same current-risethreshold is employed, changes in the voltage of the power supply 2 maycause the aligned position to be determined at an earlier point (if thevoltage increases) or at a later point (if the voltage decreases).Accordingly, the controller 13 may adjust the rise-time threshold inresponse to changes in the voltage of the power supply 2. In particular,the controller 13 may decrease the rise-time threshold in response to anincrease in the phase voltage, and vice versa. As a result, the alignedposition may be determined with improved accuracy over a range ofvoltages.

In addition to adjusting the rise-time threshold, the controller 13 mayalso adjust the freewheel period in response to changes in the voltageof the power supply 2. For example, if the voltage of the power supply 2decreases, the phase current will rise at a slower rate duringexcitation and thus the frequency of current chopping will decrease. Tocompensate for this, a shorter freewheel period may be employed. Moregenerally, the controller 13 may adjust the current limit and/or thefreewheel period in response to changes in rotor speed and/or supplyvoltage so as to better shape the phase-current waveform, therebyincreasing the power and/or efficiency of the motor system 1.

The controller 13 may also employ a speed threshold that depends on thevoltage of the power supply 2. When employing the first sensorlessscheme, the accuracy of the aligned position depends on the frequency ofcurrent chopping. As the voltage of the power supply 2 decreases, thephase current rises at a slower rate and thus the frequency of currentchopping decreases. At relatively low speeds, the frequency of currentchopping is relatively high and thus a decrease in the voltage of thepower supply 2 is unlikely to influence greatly the accuracy of thealigned position. However, at relatively high speeds where the frequencyof current chopping is relatively low, a decrease in the voltage of thepower supply may adversely affect the accuracy of the aligned position.Accordingly, for a lower supply voltage, it may be desirable to switchto the second sensorless scheme at a lower speed. When employing thesecond sensorless scheme, or rather the particular embodimentillustrated in FIGS. 6 and 7, the position of the rotor 5 can bedetermined during phase excitation only. The controller 13 thereforecommutates the phase winding 7 in advance of each aligned position so asto ensure that the phase winding 7 is excited as the rotor 5 passesthrough the aligned position. However, the controller 13 freewheels thephase winding 7 whenever current in the phase winding 7 exceeds thecurrent limit. It is therefore important that the phase current does notexceed the current limit before the rotor 5 reaches the alignedposition. The speed threshold and the advance period are thereforechosen such that, for a nominal supply voltage, the phase current doesnot exceed the current limit until after the rotor 5 has passed throughthe aligned position. If, however, the voltage of the power supply 2were to increase, the phase current would rise at a faster rate and thusthe current limit would be reached at an earlier point in time.Conceivably, the phase current may exceed the current limit before therotor 5 has reached the aligned position. Accordingly, for a highersupply voltage, it may be desirable to switch to the second sensorlessscheme at a higher speed, where the magnitude of the back EMF will behigher. Both the sensorless schemes can therefore benefit from a speedthreshold that depends on the magnitude of the supply voltage.Accordingly, the controller 13 may employ a speed threshold that dependson the magnitude of the supply voltage. More particularly, thecontroller 13 may employ a lower speed threshold for a lower supplyvoltage.

The first sensorless scheme described above uses the current-rise periodto determine the position of the rotor 5. However, the applicant hasfound that the magnitude of the phase current at the end of eachfreewheel period may also be used to determine the position of the rotor5. As noted above, the controller 13 freewheels the phase winding 7whenever current in the phase winding 7 exceeds a current limit. Thecontroller 13 freewheels the phase winding 7 for a predeterminedfreewheel period, during which time current in the phase winding 7decays. During each freewheel period, the back EMF induced in the phasewinding 7 acts in opposition to the direction of current in the phasewinding 7. The rate at which the phase current decays therefore dependson the magnitude of the back EMF. Consequently, the magnitude of thephase current at the end of each freewheel period depends on themagnitude of back EMF in the phase winding 7. The magnitude of the backEMF induced in the phase winding 7 depends on, among other things, theangular position of the rotor 5. Accordingly, the magnitude of the phasecurrent at the end of each freewheel period may be used to determine theposition of the rotor 5.

The waveform of the back EMF is typically sinusoidal (as illustrated inFIGS. 4 and 5) or trapezoidal, with zero-crossings in the back EMFoccurring at aligned positions of the rotor 5. Consequently, as therotor 5 approaches an aligned position, the magnitude of the back EMFdecreases and thus the magnitude of the phase current at the end of eachfreewheel period increases. This behaviour may then be exploited by thecontroller 13 in order to determine the position of the rotor 5. Inparticular, the controller 13 may measure the magnitude of the phasecurrent at the end of each freewheel period and compare this against acurrent threshold. When the phase current exceeds the current threshold,the controller 13 determines that the rotor 5 is at an aligned position.

FIG. 13 illustrates waveforms for the phase current, the on/off signalfor power switch Q1, and the back EMF over one electrical half-cyclewhen employing this alternative method of implementing the firstsensorless scheme. The waveforms correspond to those of FIG. 4, with theexception that the position of the rotor 5 is determined using themagnitude of the phase current at the end of each freewheel periodrather than the current-rise period.

When the current-rise period is used to determine the rotor position,the value chosen for the current-rise threshold influences the accuracyof the aligned position determined by the controller 13. Likewise, whenthe magnitude of the phase current at the end of the freewheel period isused to determine the rotor position, the value chosen for the currentthreshold influences the accuracy of the determined aligned position. Inparticular, if the current threshold is set too low, the controller 13is likely to determine an aligned position for the rotor 5 at an earlierpoint. Conversely, if the current threshold is set too high, thecontroller 13 is likely to determine an aligned position for the rotor 5at a later point.

As the rotor speed increases, the frequency of current choppingdecreases. Additionally, the magnitude of the back EMF increases andthus the phase current decays at a faster rate during each freewheelperiod. Accordingly, when the rotor 5 is at or near the alignedposition, the magnitude of the phase current at the end of the freewheelperiod is likely to be lower at higher rotor speeds. This can be seen,for example, in FIGS. 4 and 5 where the rotor speed is respectivelylower and higher. If the same current threshold were employedirrespective of rotor speed, the controller 13 would most likelydetermine that the rotor 5 is at the aligned position at an earlierpoint when operating at lower rotor speeds and a later point whenoperating at higher rotor speeds. Accordingly, in order to improve theaccuracy of the determined aligned position, the controller 13 mayemploy a current threshold that varies with rotor speed. In particular,the controller 13 may employ a current threshold that decreases withincreasing rotor speed. Consequently, at lower speeds, where thefrequency of current chopping is relatively high, a higher currentthreshold may be used. Conversely, at higher speeds, where the frequencyof current chopping is relatively low, a lower current threshold may beused. As a result, the aligned position may be determined with improvedaccuracy over a range of rotor speeds.

Two different methods are therefore available for implementing the firstsensorless scheme. In the first, the current-rise period is used todetermine the position of the rotor. In the second, the magnitude of thephase current at the end of the freewheel period is used to determinethe rotor position. Accordingly, in a more general sense, the firstsensorless scheme may be said to comprise sequentially exciting andfreewheeling the phase winding. The winding is freewheeled for apredetermined freewheel period in response to current in the windingexceeding a current limit. A parameter is then measured that correspondsto either the current-rise period or the magnitude of the phase currentat the end of the freewheel period. The measured parameter is thencompared against a threshold, and the rotor is determined to be at analigned position when the measured parameter is less than the threshold(e.g. when the current-rise period is less than the rise-time threshold)or greater than the threshold (e.g. when the magnitude of phase currentis greater than the current threshold).

The first sensorless scheme exploits the finding that the current-riseperiod decreases and the magnitude of the phase current at the end ofthe freewheel period increases as the rotor 5 approaches the alignedposition. The applicant has also found that the current-rise period isat a maximum and the magnitude of the phase current at the end of thefreewheel period is at a minimum when the rotor 5 is at or near themidpoint between the unaligned and aligned positions. Accordingly,rather than determining that the rotor 5 is at the aligned position whenthe current-rise period is less than a rise-time threshold or when thephase current at the end of the freewheel period is greater than acurrent threshold, the controller 13 may instead determine that therotor 5 is at the midpoint position when the current-rise period isgreater than a rise-time threshold or when the phase current at the endof the freewheel period is less than a current threshold. Accordingly,in a more general sense, the controller 13 may be said to determine thatthe rotor 5 is at a predetermined position when the measured parameteris less than or greater than a threshold.

Through appropriate selection of the rise-time threshold or the currentthreshold, any predetermined position for the rotor 5 may be determinedby the controller 13. For example, let us say that the controller 13determines that the rotor 5 is at a particular position when thecurrent-rise period is less than the rise-time threshold. By increasingthe rise-time threshold, the rotor position will be determined at anearlier point. Conversely, by decreasing the rise-time threshold, therotor position will be determined at a later point. The value of therise-time threshold or the current threshold may therefore be definedsuch that the compare operation (i.e. is the measured parameter lessthan or greater than the threshold) is satisfied for a particularpredetermined position of the rotor 5. Moreover, the value of thethreshold may be defined so as to control the commutation point for thephase winding 7. For example, let us say that the controller 13commutates the phase winding 7 immediately on determining that thecurrent-rise period is less than the rise-time threshold. Moreover, letus say that the value of the rise-time threshold is defined initiallysuch that current-rise period is less than the rise-time threshold whenthe rotor 5 is at the aligned position. By increasing the value of therise-time threshold, the current-rise period will be less than therise-time threshold at an earlier point and thus the controller 13 willcommutate the phase winding 7 before the aligned position, i.e. advancedcommutation. Conversely, by decreasing the value of the rise-timethreshold, the current-rise period will be less than the rise-timethreshold at later point and thus the controller 13 will commutate thephase winding 7 after the aligned position, i.e. retarded commutation.Accordingly, advanced, synchronous or retarded commutation may beachieved through appropriate selection of the rise-time threshold or thecurrent threshold. This then has the advantage that the commutationpoint may be set without the need to calculate the commutation period,T_COM, or employ a dedicated timer to measure the commutation period.

Two different methods have thus far been described for implementing thefirst sensorless scheme. In each, the controller 13 excites the phasewinding 7 until the phase current exceeds a current limit, in responseto which the controller 13 freewheels the phase winding 7 for apredetermined freewheel period. Two further methods for implementing thefirst sensorless scheme will now be described. In a third method, thecontroller 13 abandons the use of a predetermined freewheel period andinstead employs an upper current limit and a lower current limit. Thecontroller 13 then excites the phase winding 7 until the phase currentrises to the upper current limit, at which point the controller 13freewheels the phase winding 7. Freewheeling then continues until thephase current decays to the lower current limit, at which point thecontroller 13 again excites the phase winding 7. The controller 13 thenmeasures the time taken for the phase current to rise from the lowercurrent limit to the upper current limit or fall from the upper currentlimit to the lower current limit. As noted above, the magnitude of theback EMF in the phase winding 7 influences the rate at which the phasecurrent rises during excitation and falls during freewheeling.Accordingly, the time taken for the phase current to rise to the uppercurrent limit or fall to the lower current limit will depend on theangular position of the rotor 5. The controller 13 then compares themeasured time against a threshold and determines that the rotor 5 is ata predetermined position when the measured time is less than or greaterthan the threshold. For example, the controller 13 may determine thatthe rotor 5 is at an aligned position when the measured time is lessthan the threshold. In a fourth method, the controller 13 initiallyexcites the phase winding 7 until the phase current exceeds the currentlimit. The controller 13 then freewheels the phase winding 7 for apredetermined freewheel period or until the phase current falls to alower current limit. At the end of freewheeling, the controller 13 againexcites the phase winding 7. However, rather than exciting the phasewinding 7 until such time as the phase current exceeds the currentlimit, the controller 13 instead excites the phase winding 7 for apredetermined excitation period. At the end of the excitation period,the controller 13 measures the magnitude of the phase current andcompares this against a threshold. Since the magnitude of the back EMFin the phase winding 7 influences the rate at which the phase currentrises during excitation, the magnitude of the phase current at the endof the excitation period will depend on the angular position of therotor 5. The controller 13 then determines that the rotor 5 is at apredetermined position when the measured phase current is less than orgreater than the threshold. For example, the controller 13 may determinethat the rotor 5 is at an aligned position when the measured phasecurrent is greater than the threshold. A disadvantage of this fourthmethod is that the magnitude of the phase current at the end of eachexcitation period is poorly controlled. As a result, the phase currentcould potentially rise to a level that damages components of the motorsystem 1. However, this problem may be mitigated by having a relativelyshort excitation period.

Several methods are therefore available for implementing the firstsensorless scheme. Irrespective of the method, the first sensorlessscheme involves sequentially exciting and freewheeling the phase winding7. A parameter is then measured at the start or end of freewheeling.This parameter depends on the rate of change of current in the phasewinding 7 during excitation or freewheeling. For example, the parametermay be the magnitude of the phase current at the start or end offreewheeling, or the parameter may be the time required for the phasecurrent to rise to an upper current limit during excitation or fall to alower current limit during freewheeling.

The second sensorless scheme generates an edge in the BEMF signalwhenever the voltages of the first and third signals correspond. In theembodiment described above, the two signals are scaled such that thevoltages correspond whenever there is a zero-crossing in the back EMF,i.e. when the rotor 5 is at an aligned position. However, the signalsmay be scaled such that the voltages correspond at a different point inthe back EMF waveform and thus at a different rotor position. Forexample, in the example illustrated in FIG. 7, increasing the voltage ofthe first signal will cause the falling edge to be generated at anearlier point. Conversely, decreasing the voltage of the first signalwill cause the falling edge to be generated at a later point.Consequently, through appropriate scaling of the two signals, thevoltages of the two signals may be made to correspond when the rotor 5is at a particular predetermined position.

The second sensorless scheme assumes that the resistive term,i_(ph)R_(ph), is relatively small and may be ignored. If need be,however, the resistive term may be taken into account. For example, theback EMF sensor 12 may comprise an amplifier or other hardware forscaling the signal output by current sensor 11 to generate a fourthsignal having a voltage that is proportional to i_(ph)R_(ph). The backEMF sensor 12 may further comprise a summing amplifier or other hardwarethat sums the voltages of the third signal (L_(ph)di_(ph)/dt) and thefourth signal (i_(ph)R_(ph)) to generate a fifth signal having a voltagethat is proportional to i_(ph)R_(ph)+L_(ph)di_(ph)/dt. The comparator 19then compares the voltages of the first signal (V_(ph)) and the fifthsignal (i_(ph)R_(ph)+L_(ph)di_(ph)/dt), and an edge is generated in theoutput signal whenever V_(ph)=i_(ph)R_(ph)+L_(ph).di_(ph)/dt.Accordingly, in a more general sense, the second sensorless scheme maybe said to comprise generating a signal having a voltage that isproportional to V_(ph), and generating a further signal having a voltagethat depends on di_(ph)/dt. The resistive term, i_(ph)R_(ph), may beignored, in which case the voltage of the further signal is proportionalto L_(ph).di_(ph)/dt. Alternatively, the resistive term may be takeninto account, in which case the voltage of the further signal isproportional to i_(ph)R_(ph)+L_(ph)di_(ph)/dt. The voltages of the twosignals are then compared and a predetermined position of the rotor isdetermined when the two voltages correspond.

In the embodiments described above, the controller 13 generates threecontrol signals (DIRT, DIR2 and FW#) for controlling the power switchesQ1-Q4 of the inverter 9. It will be appreciated that other schemes maybe employed for controlling the power switches Q1-Q4. By way of example,the processor of the controller 13 may generate a switching signal S1-S4for each of the four power switches Q1-Q4. Hardware internal or externalto the controller 13 may then generate a current-limit signal that islogically high whenever the phase current exceeds the current limit. Thecurrent-limit signal takes precedence over the switching signals suchthat, irrespective of the states of switching signals S1-S4, thehigh-side switches Q1,Q3 are opened in response to a logically highcurrent-limit signal. Employing hardware to generate the current-limitsignal has the advantage that the control system 4 is able to respondrelatively quickly to a current-limit event.

In the embodiments described above, freewheeling involves opening thehigh-side switches Q1,Q3 and allowing current in the phase winding 7 tore-circulate around the low-side loop of the inverter 9. Conceivably,freewheeling might instead involve opening the low-side switches Q2,Q4and allowing current to re-circulate around the high-side loop of theinverter 9. Accordingly, in a more general sense, freewheeling should beunderstood to mean that zero volts are applied to the phase winding 7.In the embodiment illustrated in FIG. 9, freewheeling around thehigh-side loop of the inverter 9 is undesirable. This is because thesense resistors R1,R2 of the current sensor 11 must then be located onthe upper legs of the inverter 9. As a result, the voltage across eachsense resistor R1,R2 will be floating, making measurement of the phasecurrent difficult.

Reference has thus far been made to a motor system 1 having asingle-phase, four-pole motor 3. However, the control system 4 mightequally be used to drive a motor 3 having a fewer or greater number ofpoles. A single-phase motor 3 has the advantage that a relatively simpleand thus cheap control system 4 may be employed to control the motor 3.Existing sensorless schemes for controlling a single-phase motorgenerally suspend excitation at points in the electrical cycle for whichzero-crossings in back EMF are expected. As a result, the electricalpower driven into the motor is significantly reduced. Additionally, theefficiency of the motor may be reduced and/or the torque ripple may beincreased. In contrast, the sensorless schemes described above arecapable of sensing the position of the rotor whilst exciting the phasewinding. As a result, either scheme may be used to control asingle-phase motor without adversely affecting the electrical power,efficiency or torque ripple.

Although the sensorless schemes described above have particular benefitswhen employed in a single-phase motor, either scheme may be employed ina multi-phase motor. For a multi-phase motor, there will be periodswhere a particular phase winding is not excited. Accordingly, where thesensorless scheme relies on phase excitation in order to sense theposition of the rotor, it will be necessary to switch current sensingfrom one phase winding to another.

1. A method of controlling a brushless permanent-magnet motor, themethod comprising: generating a first signal having a voltage that isproportional to a voltage across a winding of the motor; generating asecond signal having a voltage that is proportional to a current in thewinding; differentiating the second signal to generate a third signal;comparing the voltages of the first signal and the third signal;generating an output signal in response to the comparison, wherein anedge is generated in the output signal when the voltages of the firstsignal and the third signal correspond; and commutating the winding attimes relative to edges in the output signal.
 2. The method of claim 1,wherein the voltages of the first signal and the third signal correspondin response to a zero-crossing in back EMF induced in the winding. 3.The method of claim 1, wherein the voltages of the first signal and thethird signal correspond in response to commutating the winding, and themethod comprises ignoring edges generated in response to commutating thewinding.
 4. The method of claim 1, wherein the method comprises excitingand freewheeling the winding, commutating the winding at times relativeto edges generated during excitation, and ignoring edges generatedduring freewheeling.
 5. The method of claim 1, wherein the methodcomprises exciting and freewheeling the winding, and generating thefirst and second signals during both excitation and freewheeling.
 6. Themethod of claim 1, wherein the method comprises exciting the windingwith an excitation voltage, and generating the first signal comprisesgenerating a signal having a voltage that is proportional to theexcitation voltage.
 7. A control system for a brushless permanent-magnetmotor, the control system performing a method comprising: generating afirst signal having a voltage that is proportional to a voltage across awinding of the motor; generating a second signal having a voltage thatis proportional to a current in the winding; differentiating the secondsignal to generate a third signal; comparing the voltages of the firstsignal and the third signal; generating an output signal in response tothe comparison, wherein an edge is generated in the output signal whenthe voltages of the first signal and the third signal correspond; andcommutating the winding at times relative to edges in the output signal.8. A control system for a brushless permanent-magnet motor, the controlsystem comprising: a first sensor generating a first signal having avoltage that is proportional to a voltage across a winding of the motor;a second sensor generating a second signal having a voltage that isproportional to a current in the winding; a differentiatordifferentiating the second signal and generating in response a thirdsignal; a comparator comparing the voltages of the first signal and thethird signal and generating in response an output signal, wherein anedge is generated in the output signal when the voltages of the firstsignal and the third signal correspond; and a controller generating oneor more control signals for commutating the winding at times relative toedges in the output signal.
 9. The control system of claim 8, whereinthe voltages of the first signal and the third signal correspond inresponse to a zero-crossing in back EMF induced in the winding.
 10. Thecontrol system of claim 8, wherein the voltages of the first signal andthe third signal correspond in response to commutating the winding, andthe controller ignores edges generated in response to commutating thewinding.
 11. The control system of claim 8, wherein the controllergenerates control signals for exciting and freewheeling the winding, andthe controller generates control signals for commutating the winding inresponse to edges generated during excitation and ignores edgesgenerated during freewheeling.
 12. The control system of claim 8,wherein the control system comprises an inverter to which the winding iscoupled, and the first sensor comprises a single potential dividerlocated across the inverter.
 13. The control system of claim 8, whereinthe control system comprises an inverter to which the winding iscoupled, the first sensor comprises a pair of potential dividers locatedon opposite sides of the winding, and the second sensor comprises one ofa current transducer, a current transformer and a pair of senseresistors located on opposite legs of the inverter.
 14. A motor systemcomprising a brushless permanent-magnet motor and a control system asclaimed in any one of claim 7 or 8, wherein the motor comprises a singlephase winding.